This invention pertains generally to microwave devices, and more particularly to non-reciprocal microwave devices, such as isolators and circulators. Non-reciprocal microwave devices are based on electrically insulating magnetic materials, such as the materials generally known as xe2x80x9cferritesxe2x80x9d. Their performance can be characterized by the ratio fmax/fmin, where fmin and fmax are defined as the edges of the frequency band in which the devices have acceptable operating characteristics (typically less than 1 dB insertion loss and more than 15 dB isolation). For the most advanced isolators and circulators available today this ratio is approximately 3:1. The present invention shows how the broadband performance can be improved substantially.
For both isolators and circulators, the bandwidth that has been achieved in practice is generally much smaller than that predicted by the design theories that have been developed for these devices. One reason for the failure of these theories to account satisfactorily for the observed performance is that they assume that the microwave ferrite is in a very uniform magnetic bias field (internal field). Such a very uniform field is difficult to achieve in practice, and is not usually realized in typical isolators and circulators. Another reason is that the theories make no allowance for the excess low-field, low-frequency loss observed in these devices.
The theoretical analysis of ferrite microwave devices is generally based on a xe2x80x9cconstitutivexe2x80x9d equation, expressing the relation between rf magnetic flux b vector and rf magnetic field vector h by a tensor equation of the form
b=xcexc0h.xe2x80x83xe2x80x83(1)
Here xcexc0 is the permeability of vacuum and  the permeability tensor (permeability relative to vacuum)                               μ          ↔                =                              [                                                            μ                                                                                            -                      j                                        ⁢                                          xe2x80x83                                        ⁢                    κ                                                                    0                                                                                                  j                    ⁢                                          xe2x80x83                                        ⁢                    κ                                                                    μ                                                  0                                                                              0                                                  0                                                                      μ                    z                                                                        ]                    .                                    (        2        )            
Here the dc bias field is assumed to be applied in the z-direction and a time dependence proportional to exp(jxcfx89t) is implied, where xcfx89=2xcfx80f and f is the signal frequency. The tensor components xcexc and xcexa can be calculated from the gyromagnetic equation of motion for the magnetization vector, with the result
xcexcxe2x88x92xcexa=1+fM/(fHxe2x88x92f)
xcexc+xcexa=1+fM/(fH+f)
fM=xcexc0xcex3Ms, fH=xcexc0xcex3Hintxe2x80x83xe2x80x83(3)
where xcex3 is the gyromagnetic ratio, Ms is the saturation magnetization and Hint is the internal magnetic field. Losses can be taken into account by assigning an imaginary part jxcex1Gf to the resonance frequency fH, xcex1G being the so-called xe2x80x9cGilbert damping parameterxe2x80x9d.
The propagation of electromagnetic waves in an unbounded ferrite medium can easily be analyzed on the basis of Eqs. (1-3). Such an analysis shows that, in general, two types of waves or xe2x80x9cwave modesxe2x80x9d exist for any propagation direction. For propagation orthogonal to the bias field, one of the modes is characterized by an rf magnetic field in the z-direction and an effective permeability equal to xcexcz, whereas the other mode is characterized by an rf magnetic field having x- and y-components, and an effective permeability given by
xcexce=(xcexc2xe2x88x92xcexa2)/xcexc.xe2x80x83xe2x80x83(4)
In the literature this scalar permeability is generally referred to as the xe2x80x9ceffectivexe2x80x9d permeability, and this custom is therefore also adopted in the present patent application. It plays an important role in the analysis of edge-mode isolators and stripline/microstrip circulators. It should be kept in mind, however, that the expression given in Eq. (4) generally does not represent an effective permeability for a guided wave in a ferrite substrate. From (3) and (4), the tensor components and the effective permeability can readily be shown to be
xe2x80x83xcexc=[fH(fH+fM)xe2x88x92f2]/(fH2xe2x88x92fH2)
xcexa=xe2x88x92fMf/(fHxe2x88x92f2)
xcexce=[(fH+fM)2xe2x88x92f2]/[fH(fH+fM)xe2x88x92f2]xe2x80x83xe2x80x83(5)
In the analysis of broadband isolators and circulators, the case in which fH is very small compared to fM and f is of special significance. If damping is neglected, Eq. (5) easily reduces to
xcexc=1
xcexa=fM/f
xcexce=1xe2x88x92(fM/f)2xe2x80x83xe2x80x83(6)
under these conditions, which implies that xcexce is negative for frequencies less than fM.
The most successful broadband isolators currently available are based on the edge-mode configuration, described in a paper entitled xe2x80x9cReciprocal and Nonreciprocal Modes of Propagation in Ferrite Stripline and =Microstrip Devicesxe2x80x9d by M. E. Hines [IEEE Trans. MTT-19, pp. 442-451, 1971]. These devices typically include a stripline or microstrip line on a ferrite substrate, which is magnetized normal to its plane. A sheet of resistive material with a predetermined surface resistance is located along the side of the strip conductor, in a plane orthogonal to the strip conductor. Hines describes a simple approximate analysis, which applies to this structure if the strip conductor is much wider than the substrate thickness. Under these conditions, the actual boundary conditions that exist at the edge of the strip conductor can be replaced by so-called xe2x80x9cmagnetic wallxe2x80x9d boundary conditions by way of approximation. This approximate procedure may be justified by the observation that any electric current in the strip conductor can not flow orthogonal to the edge, and hence cannot induce a magnetic field component parallel to the strip conductor. For magnetic wall boundary conditions, the field equations can be solved exactly and simply, as shown by Hines. His analysis shows that the fundamental mode of this structure, with the resistive plane removed, consists of a wave that propagates parallel to the strip conductor and varies exponentially in the transverse direction. Thus the energy carried by the wave is displaced predominantly to one side of the strip conductor. The dispersion relation for these waves can be characterized by a scalar permeability, which turns out to be equal to the diagonal component of the permeability tensor, not the effective permeability of Eq. (4).
The effect of the resistive layer on the propagation characteristics of the edge-mode isolator has also been analyzed by Hines. Because of the field displacement effect mentioned above, the attenuation depends on the direction of propagation. The presence of a resistive layer with a given surface resistance (ohm per square) can be taken into account by imposing the appropriate transverse impedance condition on the rf field. The resultant characteristic equation for the complex propagation constant xcex2 as a function of frequency xcfx89 can be expressed as F (xcex2,xcfx89)=0, where F is a relatively simple transcendental function that depends on all relevant device parameters. Solutions for the propagation constant can be constructed by Newton""s method for both directions of propagation, and for the dominant mode as well as any higher-order mode. Hines has reported the results of such calculation, taking only the losses due to the resistive layer into account and assuming a homogeneous bias field. The difference in the attenuation constants can be very large when the strip conductor is sufficiently wide.
Hines has also pointed out that, on the basis of the theory he developed, one might expect the edge-mode circulator to work over a virtually unlimited bandwidth if the ferrite is biased to saturation and the internal magnetic bias field is suitably small. In his experiments he obtained a frequency ratio fmax/fmin of about 2:1, which was considered very good at the time. Later investigators have improved the bandwidth somewhat and have achieved fmax/fmin of approx. 3:1. The discrepancy between the theoretically expected bandwidth and that obtained in practice has traditionally been attributed to xe2x80x9clow-field lossxe2x80x9d, but the exact nature of this loss has remained mysterious. It is well known that unmagnetized and partially magnetized ferrite materials are very lossy when the signal frequency f is less than the characteristic frequency fM defined in Eq. (3), i. e. for
fxe2x89xa6fM.xe2x80x83xe2x80x83(7)
This behavior can be explained by noting that, in magnetic materials that contain domains of opposite polarity, an unusual type of ferromagnetic resonance can occur. But this mechanism does not apply to a magnetically saturated ferrite, and hence does not actually explain the low-field loss observed in the edge-mode isolators.
Broadband circulators are usually based on the stripline or the microstrip configuration. The stripline version typically includes a symmetric three-way junction of strip conductors connected to a central metal disk and sandwiched between ferrite substrates or substrates that are part ferrite part dielectric. In either case, the substrates are magnetized orthogonal to their plane. The volume underneath and above the central metal disk is generally occupied by ferrite, but the ferrite may extend further out from the junction center. As first pointed out in a paper entitled xe2x80x9cOn Stripline Y-Circulation at UHFxe2x80x9d by H. Bosma [IEEE Trans, MTT-12, pp. 61-72, January 1964], this structure can be conveniently analyzed by introducing a Green""s function G(r,xcfx86;R,xcfx86xe2x80x2) that relates the axial component of the electrical field ez(r,xcfx86) at an arbitrary point (r,xcfx86) within the ferrite disc to the circumferential component of magnetic field hxcfx86(R,xcfx86xe2x80x2) at the periphery (R,xcfx86xe2x80x2) of the disk. The Green""s function is derived from Maxwell""s equations for the region occupied by the ferrite, in which the rf permeability has the form given in Eqs. (1) and (2). The scattering matrix of the circulator can then be calculated, using the assumption that the circumferential component of magnetic field is zero along the periphery of the disk, except where it is connected to the strip conductors. In the latter regions the rf magnetic field is determined by the incoming and outgoing electromagnetic waves.
The broadband circulator analysis based on Bosma""s approach was further developed in the paper entitled xe2x80x9cWideband Operation of Microstrip Circulatorsxe2x80x9d by Y. S. Wu and F. J. Rosenbaum [IEEE Trans. MTT-22, pp. 849-856, October 1974] and the paper entitled xe2x80x9cThe Frequency Behavior of Stripline Circulator Junctionsxe2x80x9d by S. Ayter and Y. Ayasli [IEEE Trans. MTT-26, pp. 197-202, March 1978]. The theory was at first developed only for the frequency range in which xcexce is positive. The fmax/fmin ratio for circulators, obtainable by this approach, is approx. 2:1. Circulator operation in the frequency range in which xcexce is negative was apparently considered impossible, because the Green""s function and the scattering matrix derived from it are represented by algebraic expressions that involve {square root over (xcexce)}, and hence appear to become very singular in the limit of very small xcexce. It is now known, however, that the apparent singularity of the Green""s function is quite harmless, because the vanishing denominators are all canceled by vanishing numerators, as shown in the paper entitled xe2x80x9cBroadband Stripline Circulators Based on YIG and Li-Ferrite Single Crystalsxe2x80x9d by E. Schloemann and R. E. Blight [IEEE Trans. NM-34, pp. 1394-1400, December 1986]. Thus the theoretical expressions derived by Bosma, and Wu/Rosenbaum remain valid when xcexce approaches zero and then becomes negative, except that the Bessel functions that occur in these expressions must now be interpreted as functions of a complex variable. For the lossless case this means that, for each order n, the Bessel function Jn is replaced by the modified Bessel function In in the manner detailed by Schloemann and Blight. The physical significance of resonant modes for xcexce less than 0 is that for these modes the excitation is large near the surface and decays toward the interior, whereas for xcexce greater than 0 the modes have an oscillatory behavior in the radial direction.
With suitably chosen design parameters (such as disk diameter, saturation magnetization, bias field, and the characteristic impedance of the transmission lines connected to the junction) the theoretically expected performance of broadband circulators according to the revised theory described by Schloemann/Blight and in the paper entitled xe2x80x9cCirculators for Microwave and Millimeter Wave Circuitsxe2x80x9d by Schloemann [Proc. IEEE, Vol. 76, pp. 188-200, February 1988] is much better than that to be expected according to the earlier calculations of Bosma, Wu/Rosenberg and Ayter/Ayasli. This may be seen from FIG. 2 of the Schloemann/Blight publication and FIG. 5 of the last quoted Schloemann publication. These figures show that the analysed circulators, when connected to transmission lines having a suitably low characteristic impedance, would have acceptable performance over a band stretching from about 0.5 GHz to 10 GHz (for the Schloemann/Blight reference) or 17 GHz (for the Schloemann reference). However, as in the case of broadband edge-mode isolators, this ideal behavior again is not observed in practical devices.
In the experimental work reported in the preceeding two references, a concerted effort was made to generate a homogeneous internal magnetic bias field, by means of hemispherical pole caps positioned outside the stripline device. The results showed that low-loss circulator operation was indeed achievable in the frequency range 0.5 fM less than f less than 2fM, but that for f less than 0.5fM some additional losses were present that could not readily be explained.
An alternative approach toward improving the broadband performance of circulators is to position a ring of a secondary ferrite having a lower saturation magnetization around the primary ferrite, which is at the junction center. This approach, which may be used for microstrip as well as for stripline circulators, has been described by M. G. Matthew and T. J. Weisz in the U.S. patents entitled xe2x80x9cMicrowave Transmission Devices Comprising Gyromagnetic Material Having Smoothly Varying Saturation Magnetizationxe2x80x9d, [U.S. Pat. No. 4,390,853] and xe2x80x9cMicrowave Transmission Devices Having Gyromagnetic Materials Having Different Saturation Magnetizationsxe2x80x9d[U.S. Pat. Nos. 4,496,915], and by R. Blight and E. Schloemann in the paper entitled xe2x80x9cA Compact Broadband Microstrip Circulator for Phased Array Antenna Modulesxe2x80x9d [IEEE MTT-S Digest, pp. 1389-1392, 1992]. It has led to the development of useful broad band chculators with an fmax/fmin ratio of about 3.
The performance of broadband non-reciprocal microwave devices (isolators and circulators), expressed as the frequency ratio fmax/fmin, is presently limited by the combination of inhomogeneity of the internal bias field and a universal low-field, low-frequency loss component. Unlike other types of low-field loss, this component occurs in fully saturated magnetic matials, and generally increases the insertion loss of ferrite microwave devices below a characteristic frequency. Formerly unexplained, this loss is now interpreted as arising from the excitation of magnetostatic surface waves (MSSWs) at the perimeter of the ferrite disc, as discussed in more detail in the Detailed Description of the Invention. According to the present invention, the MSSW-related loss can be shifted out of the desired performance band of the device by maintaining a homogeneous internal magnetic bias field, and by using a multiplicity of magnetic materials, arranged in a sequence, such that the material having the highest magnetization is at or near the center of the device and materials with progressively lower magnetizations are further away from the center.
In a first preferred embodiment, these materials are placed inside the microwave transmission structure, and high-permeability pole pieces are placed adjacent to the thin conductive envelop of the microwave transmission structure as part of the magnetic bias circuit. In a second preferred embodiment, the magnetic materials having different saturation magnetizations are placed inside the microwave transmission structure, and also outside the microwave transmission structure. In this embodiment, composite pole shoes of the magnetic materials having different saturation magnetizations are placed between the thin conductive envelop of the microwave transmission structure and the high-permeability pole pieces, as part of the magnetic bias circuit. The purpose of the composite magnetic pole shoes is to improve the homogeneity of the internal magnetic bias field, to which the pieces of magnetic material inside the microwave transmission line are exposed. This bias field is adjusted to have a suitably small value.
In accordance with the present invention, a broadband non-reciprocal microwave device includes a multi-port transmission line structure that contains a multiplicity of magnetic materials interior to its conductive envelop; arranged in a sequence, such that the material having the highest magnetization is at or near the center of the device and materials with progressively lower magnetizations are further away from the center. The microwave device further includes a magnetic bias circuit designed to generate a homogeneous internal magnetic field inside each of the magnetic materials, this magnetic field being very small compared to the smallest saturation magnetization.
In accordance with a further aspect of the present invention, a broadband non-reciprocal microwave device includes facilities for providing isolator action in a two-port transmission line that contains at least two magnetic materials interior to its conductive envelop. The magnetic materials are arranged in a sequence, such that the material having the highest magnetization is at or near the center of the device, and materials with progressively lower magnetizations are further away from the center. The facilities for providing isolator action further include a resistive sheet of material with a predetermined surface resistance disposed along the microwave transmission line in an off-center position, and a magnetic bias circuit designed to generate a uniform internal magnetic field inside each of the magnetic materials, this magnetic field being very small compared to the smallest saturation magnetization.
In accordance with a further aspect of the present invention, a broadband non-reciprocal microwave device includes facilities for providing circulator action in a three-port transmission line structure that contains at least two magnetic materials interior to its conductive envelop arranged in a sequence, such that the material having the highest magnetization is at or near the center of the device, and materials with progressively lower magnetizations are further away from the center. The facilities for providing circulator action further include a magnetic bias circuit designed to generate a uniform internal magnetic field inside each of the magnetic materials, this magnetic field being very small compared to the smallest saturation magnetization.